Oversampling in a combined transposer filterbank

ABSTRACT

The present invention relates to coding of audio signals, and in particular to high frequency reconstruction methods including a frequency domain harmonic transposer. A system and method for generating a high frequency component of a signal from a low frequency component of the signal is described. The system comprises an analysis filter bank ( 501 ) comprising an analysis transformation unit ( 601 ) having a frequency resolution of Δf; and an analysis window ( 611 ) having a duration of D A ; the analysis filter bank ( 501 ) being configured to provide a set of analysis subband signals from the low frequency component of the signal; a nonlinear processing unit ( 502, 650 ) configured to determine a set of synthesis subband signals based on a portion of the set of analysis subband signals, wherein the portion of the set of analysis subband signals is phase shifted by a transposition order T; and a synthesis filter bank ( 504 ) comprising a synthesis transformation unit ( 602 ) having a frequency resolution of QΔf; and a synthesis window ( 612 ) having a duration of D S ; the synthesis filter bank ( 504 ) being configured to generate the high frequency component of the signal from the set of synthesis subband signals; wherein Q is a frequency resolution factor with Q≧1 and smaller than the transposition order T; and wherein the value of the product of the frequency resolution Δf and the duration D A  of the analysis filter bank is selected based on the frequency resolution factor Q.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.13/499,893 filed Apr. 2, 2012, which is a National Phase entry of PCTPatent Application Serial No. PCT/EP2010/057156, having internationalfiling date of May 25, 2010 and entitled “Oversampling in a CombinedTransposer Filterbank” which claims priority to U.S. Provisional PatentApplication No. 61/330,786, filed May 3, 2010 and U.S. ProvisionalPatent Application No. 61/253,775, filed Oct. 21, 2009. The contents ofall of the above applications are incorporated by reference in theirentirety for all purposes.

TECHNICAL FIELD

The present invention relates to coding of audio signals, and inparticular to high frequency reconstruction methods including afrequency domain harmonic transposer.

BACKGROUND OF THE INVENTION

HFR technologies, such as the Spectral Band Replication (SBR)technology, allow to significantly improve the coding efficiency oftraditional perceptual audio codecs. In combination with MPEG-4 AdvancedAudio Coding (AAC), HFR technologies form very efficient audio codecs,which are already in use within the XM Satellite Radio system andDigital Radio Mondiale, and also standardized within 3GPP, DVD Forum andothers. The combination of AAC and SBR is called aacPlus. It is part ofthe MPEG-4 standard where it is referred to as the High Efficiency AACProfile (HE-AAC). In general, HFR technology can be combined with anyperceptual audio codec in a back and forward compatible way, thusoffering the possibility to upgrade already established broadcastingsystems like the MPEG Layer-2 used in the Eureka DAB system. HFRtransposition methods can also be combined with speech codecs to allowwide band speech at ultra low bit rates.

The basic idea behind HRF is the observation that usually a strongcorrelation between the characteristics of the high frequency range of asignal and the characteristics of the low frequency range of the samesignal is present. Thus, a good approximation for a representation ofthe original input high frequency range of a signal can be achieved by asignal transposition from the low frequency range to the high frequencyrange.

This concept of transposition was established in WO 98/57436 which isincorporated by reference, as a method to recreate a high frequency bandfrom a lower frequency band of an audio signal. A substantial saving inbit-rate can be obtained by using this concept in audio coding and/orspeech coding. In the following, reference will be made to audio coding,but it should be noted that the described methods and systems areequally applicable to speech coding and in unified speech and audiocoding (USAC).

In a HFR based audio coding system, a low bandwidth signal is presentedto a core waveform coder for encoding, and higher frequencies areregenerated at the decoder side using transposition of the low bandwidthsignal and additional side information, which is typically encoded atvery low bit-rates and which describes the target spectral shape. Forlow bit-rates, where the bandwidth of the core coded signal is narrow,it becomes increasingly important to reproduce or synthesize a highband, i.e. the high frequency range of the audio signal, withperceptually pleasant characteristics.

One of the underlying problems that exist with harmonic HFR methods arethe opposing constraints of an intended high frequency resolution inorder to get a high quality transposition for stationary sounds, and thetime response of the system for transient or percussive sounds. In otherwords, while the use of a high frequency resolution is beneficial forthe transposition of stationary signals, such high frequency resolutiontypically requires large window sizes which are detrimental when dealingwith transient portions of a signal. One approach to deal with thisproblem may be to adaptively change the windows of the transposer, e.g.by using window-switching, as a function of input signalcharacteristics. Typically long windows will be used for stationaryportions of a signal, in order to achieve high frequency resolution,while short windows will be used for transient portions of the signal,in order to implement a good transient response, i.e. a good temporalresolution, of the transposer. However, this approach has the drawbackthat signal analysis measures such as transient detection or the likehave to be incorporated into the transposition system. Such signalanalysis measures often involve a decision step, e.g. a decision on thepresence of a transient, which triggers a switching of signalprocessing. Furthermore, such measures typically affect the reliabilityof the system and they may introduce signal artifacts when switching thesignal processing, e.g. when switching between window sizes.

In order to reach improved audio quality and in order to synthesize therequired bandwidth of the high band signal, harmonic HFR methodstypically employ several orders of transposition. In order to implementa plurality of transpositions of different transposition order, priorart solutions require a plurality of filter banks either in the analysisstage or the synthesis stage or in both stages. Typically, a differentfilter bank is required for each different transposition order.Moreover, in situations where the core waveform coder operates at alower sampling rate than the sampling rate of the final output signal,there is typically an additional need to convert the core signal to thesampling rate of the output signal, and this upsampling of the coresignal is usually achieved by adding yet another filter bank. All inall, the computationally complexity increases significantly with anincreasing number of different transposition orders.

The present document addresses the aforementioned problems regarding thetransient performance of harmonic transposition and regarding thecomputational complexity. As a result, improved harmonic transpositionis achieved at a low additional complexity.

SUMMARY OF THE INVENTION

According to an aspect, a system configured to generate a high frequencycomponent of a signal from a low frequency component of the signal isdescribed. The system may comprise an analysis filter bank comprising ananalysis transformation unit having a frequency resolution of Δf. Theanalysis transformation unit may be configured to perform e.g. a FourierTransform, a Fast Fourier Transform, a Discrete Fourier Transform or aWavelet Transform. The analysis filter bank may further comprise ananalysis window having a duration of D_(A). The analysis window may havethe shape e.g. of a Gaussian window; a cosine window; a Hamming window;a Hann window; a rectangular window; a Bartlett window; or a Blackmanwindow. The analysis filter bank may be configured to provide a set ofanalysis subband signals from the low frequency component of the signal.

The system may comprise a nonlinear processing unit configured todetermine a set of synthesis subband signals based on a portion of theset of analysis subband signals, wherein the portion of the set ofanalysis subband signals is phase shifted by a transposition order T. Inparticular, the subband signals may comprise complex values and thephase shifting may comprise the multiplication of the phase of thecomplex subband values by the order T.

The system may comprise a synthesis filter bank comprising a synthesistransformation unit having a frequency resolution of QΔf. The synthesistransformation unit may be configured to perform the correspondinginverse transform to the transform performed by the analysistransformation unit.

Furthermore, the synthesis filter bank may comprise a synthesis windowhaving a duration of D_(S) and having any of the above listed shapes. Qis a frequency resolution factor with Q≧1 and smaller than thetransposition order T. In a particular embodiment, the frequencyresolution factor is selected as Q>1. The synthesis filter bank may beconfigured to generate the high frequency component of the signal fromthe set of synthesis subband signals.

Typically, the value of the product of the frequency resolution Δf andthe duration D_(A) of the analysis filter bank is selected based on thefrequency resolution factor Q. In particular, the product ΔfD_(A) may beproportional to

$\frac{1}{Q + 1}.$In an embodiment, the value of product the ΔfD_(A) is smaller or equalto

$\frac{2}{Q + 1}.$Furthermore, the product ΔfD_(A) may be greater than

$\frac{2}{T + 1}.$The value of the product ΔfD_(A) of the analysis filter bank may beequal to the value of the product QΔfD_(S) of the synthesis filter bank.By selecting the analysis and/or the synthesis filter bank according toany of the above rules, artifacts caused by harmonic transposition ontransients of the signal may be reduced or completely removed, whileallowing a reduced computational complexity of the harmonic transposer.

The system may further comprise a second nonlinear processing unitconfigured to determine a second set of synthesis subband signals fromthe set of analysis subband signals using a second transposition orderT₂; wherein the second set of synthesis subband signals is determinedbased on a portion of the set of analysis subband signals and whereinthe portion of the set of analysis subband signals is phase shifted bythe second transposition order T₂. The transposition order T and thesecond transposition order T₂ may be different. The system may furthercomprise a combining unit configured to combine the set of synthesissubband signals and the second set of synthesis subband signals; therebyyielding a combined set of synthesis subband signals as an input to thesynthesis filter bank. The combining unit may be configured to add oraverage corresponding subband signals from the set of synthesis subbandsignals and the second set of synthesis subband signals. In other words,the combining unit may be configured to superpose synthesis subbandsignals of the set of synthesis subband signals and the second set ofsynthesis subband signals corresponding to overlapping frequency ranges.

In an embodiment, the analysis filter bank may have a number K_(A) ofanalysis subbands, with K_(A)>1, where k is an analysis subband indexwith k=0, . . . , K_(A)−1. The synthesis filter bank may have a numberN_(S) of synthesis subbands, with N_(S)>0, where n is a synthesissubband index with n=0, . . . , N_(S)−1. In such cases, the nonlinearprocessing unit may be configured to determine an n^(th) synthesissubband signal of the set of synthesis subband signals from a k^(th)analysis subband signal and a (k+1)^(th) analysis subband signal of theset of analysis subband signals. In particular, the nonlinear processingunit may be configured to determine a phase of the n^(th) synthesissubband signal as the sum of a shifted phase of the k^(th) analysissubband signal and a shifted phase of the (k+1)^(th) analysis subbandsignal. Furthermore, the nonlinear processing unit may be configured todetermine a magnitude of the n^(th) synthesis subband signal as theproduct of an exponentiated magnitude of the k^(th) analysis subbandsignal and an exponentiated magnitude of the (k+1)^(th) analysis subbandsignal.

The analysis subband index k of the analysis subband signal contributingto the synthesis subband with synthesis subband index n may be given bythe integer obtained by truncating the expression

${\frac{Q}{T}n};$wherein a remainder r may be given by

${\frac{Q}{T}n} - {k.}$In such cases, the nonlinear processing unit may be configured todetermine the phase of the n^(th) synthesis subband signal as the sum ofthe phase of the k^(th) analysis subband signal multiplied by T(1−r) andthe phase of the (k+1)^(th) analysis subband signal multiplied by T(r),i.e. by performing a linear interpolation of phase. Furthermore, thenonlinear processing unit may be configured to determine the magnitudeof the n^(th) synthesis subband signal as the product of the magnitudeof the k^(th) analysis subband signal raised to the power of (1−r) andthe magnitude of the (k+1)^(th) analysis subband signal raised to thepower of r, i.e. by determining the geometrical mean of the magnitudes.

The analysis filter bank and the synthesis filter bank may be evenlystacked such that a center frequency of an analysis subband is given bykΔf and a center frequency of a synthesis subband is given by nQΔf. Inan alternative embodiment, the analysis filter bank and the synthesisfilter bank may be oddly stacked such that a center frequency of ananalysis subband is given by

$\left( {k + \frac{1}{2}} \right)\Delta\; f$and a center frequency of a synthesis subband is given by

${\left( {n + \frac{1}{2}} \right)Q\;\Delta\; f};$and the difference between the transposition order T and the resolutionfactor Q is even.

A sampling rate of the low frequency component may be f_(A). Theanalysis transformation unit may perform a discrete M pointtransformation. The analysis window may have a length of L_(A) samplesand/or the analysis window may be shifted by an analysis hop size ofΔs_(A) samples along the low frequency component. In such cases, thefrequency resolution may be given by

${{\Delta\; f} = \frac{f_{A}}{M}},$the duration of the analysis window may be given by

$D_{A} = \frac{L_{A}}{f_{A}}$and/or a physical time stride of the analysis filter bank may be givenby

${\Delta\; f_{A}} = {\frac{\Delta\; s_{A}}{f_{A}}.}$

A sampling rate of the high frequency component may be f_(S)=Qf_(A). Thesynthesis transformation unit may perform a discrete M pointtransformation, in particular it may perform the respective inversetransformation of the analysis transformation unit. The synthesis windowmay have a length of L_(S) samples and/or the synthesis window may beshifted by a synthesis hop size of Δs_(S) samples along the highfrequency component. In such cases, the frequency resolution may begiven by

${{Q\;\Delta\; f} = \frac{f_{s}}{M}},$the duration may be given by

$D_{s} = \frac{L_{s}}{f_{s}}$and/or a physical time stride of the synthesis filter bank may be givenby

${\Delta\; t_{s}} = {\frac{\Delta\; s_{s}}{f_{s}} = {\frac{\Delta\; s_{A}}{f_{A}} = {\Delta\;{t_{A}.}}}}$

According to a further aspect, a system for generating an output signalcomprising a high frequency component from an input signal comprising alow frequency component using a transposition order T is described. Thesystem may comprise an analysis window unit configured to apply ananalysis window of a length of L_(A) samples, thereby extracting a frameof the input signal. The system may comprise an analysis transformationunit of order M and having a frequency resolution Δf configured totransform the L_(A) samples into M complex coefficients. The system maycomprise a nonlinear processing unit, configured to alter the phase ofthe complex coefficients by using the transposition order T. Thealtering of the phase may comprise shifting the phase of the complexcoefficients as outlined in the present document. The system maycomprise a synthesis transformation unit of order M and having afrequency resolution QΔf, configured to transform the alteredcoefficients into M altered samples; wherein Q is a frequency resolutionfactor smaller than the transposition order T. Furthermore, the systemmay comprise a synthesis window unit configured to apply a synthesiswindow of a length of L_(S) samples to the M altered samples, therebygenerating a frame of the output signal.

M may be based on the frequency resolution factor Q. In particular, thedifference between M and the average length of the analysis window andthe synthesis window (612) may be proportional to (Q−1). In anembodiment, M is greater or equal to (QL_(A)+L_(S))/2. Furthermore, Mmay be smaller than (TL_(A)+L_(S))/2.

According to another aspect, a method for generating a high frequencycomponent of a signal from a low frequency component of the signal isdescribed. The method may comprise the step of providing a set ofanalysis subband signals from the low frequency component of the signalusing an analysis filter bank comprising an analysis transformation unithaving a frequency resolution of Δf and an analysis window having aduration of D_(A). Furthermore, the method may comprise the step ofdetermining a set of synthesis subband signals based on a portion of theset of analysis subband signals, wherein the portion of the set ofanalysis subband signals is phase shifted by a transposition order T.Eventually, the method may comprise the step of generating the highfrequency component of the signal from the set of synthesis subbandsignals using a synthesis filter bank comprising a synthesistransformation unit having a frequency resolution of QΔf and a synthesiswindow having a duration of D_(S). Q is a resolution factor with Q≧1 andsmaller than the transposition order T. The value of the product of thefrequency resolution Δf and the duration D_(A) of the analysis filterbank may be selected based on the frequency resolution factor Q.

According to a further aspect, a method for generating an output signalcomprising a high frequency component from an input signal comprising alow frequency component using a transposition order T is described. Themethod may comprise the steps of applying an analysis window of a lengthof L_(A) samples, thereby extracting a frame of the input signal; and oftransforming the frame of L_(A) samples of the input signal into Mcomplex coefficients using an analysis transformation of order M andfrequency resolution Δf. Furthermore, the method may comprise the stepof altering the phase of the complex coefficients by using thetransposition order T. The altering of the phase may be performedaccording to the methods outlined in the present document. In addition,the method may comprise the steps of transforming the alteredcoefficients into M altered samples using a synthesis transformation oforder M and of frequency resolution QΔf, wherein Q is a frequencyresolution factor smaller than the transposition order T; and ofapplying a synthesis window of a length of L_(S) samples to the Maltered samples, thereby generating a frame of the output signal. M maybe based on the frequency resolution factor Q.

According to another aspect, a method for designing a harmonictransposer configured to generate a high frequency component of a signalfrom a low frequency component of the signal is described. The methodmay comprise the step of providing an analysis filter bank comprising ananalysis transformation unit having a frequency resolution of Δf; and ananalysis window having a duration of D_(A); the analysis filter bankbeing configured to provide a set of analysis subband signals from thelow frequency component of the signal. Furthermore, the method maycomprise the step of providing a nonlinear processing unit configured todetermine a set of synthesis subband signals based on a portion of theset of analysis subband signals, wherein the portion of the set ofanalysis subband signals is phase shifted by a transposition order T. Inaddition, the method may comprise the step of providing a synthesisfilter bank comprising a synthesis transformation unit having afrequency resolution of QΔf; and a synthesis window having a duration ofD_(S); the synthesis filter bank being configured to generate the highfrequency component of the signal from the set of synthesis subbandsignals; wherein Q is a frequency resolution factor with Q≧1 and smallerthan the transposition order T. Furthermore, the method may comprise thestep of selecting the value of the product of the frequency resolutionΔf and the duration D_(A) of the analysis filter bank based on thefrequency resolution factor Q.

According to another aspect, a method for designing a transposerconfigured to generate an output signal comprising a high frequencycomponent from an input signal comprising a low frequency componentusing a transposition order T is described. The method may comprise thesteps of providing an analysis window unit configured to apply ananalysis window of a length of L_(A) samples, thereby extracting a frameof the input signal; and of providing an analysis transformation unit oforder M and having a frequency resolution Δf configured to transform theL_(A) samples into M complex coefficients. Furthermore, the method maycomprise the step of providing a nonlinear processing unit, configuredto alter the phase of the complex coefficients by using thetransposition order T. In addition, the method may comprise the steps ofproviding a synthesis transformation unit of order M and having afrequency resolution QΔf, configured to transform the alteredcoefficients into M altered samples; wherein Q is a frequency resolutionfactor smaller than the transposition order T; and of providing asynthesis window unit configured to apply a synthesis window of a lengthof L_(S) samples to the M altered samples, thereby generating a frame ofthe output signal. Eventually, the method may comprise the step ofselecting M based on the frequency resolution factor Q.

It should be noted that the methods and systems including its preferredembodiments as outlined in the present patent application may be usedstand-alone or in combination with the other methods and systemsdisclosed in this document. Furthermore, all aspects of the methods andsystems outlined in the present patent application may be arbitrarilycombined. In particular, the features of the claims may be combined withone another in an arbitrary manner.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will now be described by way of illustrativeexamples, not limiting the scope or spirit of the invention, withreference to the accompanying drawings, in which:

FIG. 1 illustrates the operation of an example single order frequencydomain (FD) harmonic transposer;

FIG. 2 illustrates the operation of an example harmonic transposer usingseveral orders;

FIG. 3 illustrates prior art operation of an example harmonic transposerusing several orders of transposition, while using a common analysisfilter bank;

FIG. 4 illustrates prior art operation of an example harmonic transposerusing several orders of transposition, while using a common synthesisfilter bank;

FIG. 5 illustrates the operation of an example harmonic transposer usingseveral orders of transposition, while using a common synthesis filterbank and a common synthesis filter bank;

FIG. 5b illustrates an example for the mapping of subband signals for amultiple transposer scheme according to FIG. 5;

FIG. 6a illustrates an example multiple transposer of order T=2, 3, 4using a common analysis filter bank and separate synthesis filter banks;

FIG. 6b illustrates an example multiple transposer of order T=2, 3, 4using a common analysis filter bank and a common synthesis filter bank;

FIG. 7 illustrates an example for the mapping of subband signals for amultiple transposer according to FIG. 6 b;

FIG. 8 illustrates a Dirac at a particular position as it appears in theanalysis and synthesis windows of a harmonic transposer;

FIG. 9 illustrates a Dirac at a different position as it appears in theanalysis and synthesis windows of a harmonic transposer; and

FIG. 10 illustrates a Dirac for the position of FIG. 9 as it will appearwhen using frequency domain oversampling.

DESCRIPTION OF PREFERRED EMBODIMENTS

The below-described embodiments are merely illustrative for theprinciples of the present invention for oversampling in a combinedtransposer filter bank. It is understood that modifications andvariations of the arrangements and the details described herein will beapparent to others skilled in the art. It is the intent, therefore, tobe limited only by the scope of the impending patent claims and not bythe specific details presented by way of description and explanation ofthe embodiments herein.

FIG. 1 illustrates the operation of a frequency domain (FD) harmonictransposer 100. In a basic form, a T^(th) order harmonic transposer is aunit that shifts all signal components H(f) of the input signal, i.e. asubband of the signal in the frequency domain, to H(Tf). I.e. thefrequency component H(f) of the input signal is shifted to a T timeshigher frequency. In order to implement such transposition in thefrequency domain, an analysis filter bank 101 transforms the inputsignal from the time-domain to the frequency domain and outputs complexsubbands or subband signals, also referred to as the analysis subbandsor analysis subband signals. The analysis filter bank typicallycomprises an analysis transform, e.g. an FFT, DFT or wavelet transform,and a sliding analysis window. The analysis subband signals aresubmitted to nonlinear processing 102 modifying the phase and/or theamplitude according to the chosen transposition order T. Typically, thenonlinear processing outputs a number of subband signals which is equalto the number of input subband signals, i.e. equal to the number ofanalysis subband signals. The modified subbands or subband signals,which are also referred to as the synthesis subbands or synthesissubband signals, are fed to a synthesis filter bank 103 which transformsthe subband signals from the frequency domain into the time domain andoutputs the transposed time domain signal. The synthesis filter bank 103typically comprises an inverse transform, e.g. an inverse FFT, inverseDFT or inverse wavelet transform, in combination with a slidingsynthesis window.

Typically, each filter bank has a physical frequency resolution Δfmeasured in Hertz and a physical time stride parameter Δt measured inseconds, wherein the physical frequency resolution Δf is usuallyassociated with the frequency resolution of the transform function andthe physical time stride parameter Δt is usually associated with thetime interval between succeeding window functions. These two parameters,i.e. the frequency resolution and the time stride, define thediscrete-time parameters of the filter bank given the chosen samplingrate. By choosing the physical time stride parameters, i.e. the timestride parameter measured in time units e.g. seconds, of the analysisand synthesis filter banks to be identical, an output signal of thetransposer 100 may be obtained which has the same sampling rate as theinput signal. Furthermore, by omitting the nonlinear processing 102 aperfect reconstruction of the input signal at the output may beachieved. This requires a careful design of the analysis and synthesisfilter banks. On the other hand, if the output sampling rate is chosento be different from the input sampling rate, a sampling rate conversionmay be obtained. This mode of operation may be necessary in the casewhere the desired bandwidth of the output signal y is larger than halfof sampling rate of the input signal x, i.e. when the desired outputbandwidth exceeds the Nyqvist frequency of the input signal.

FIG. 2 illustrates the operation of a multiple transposer or multipletransposer system 200 comprising several harmonic transposers 201-1, . .. , 201-P of different orders. The input signal which is to betransposed is passed to a bank of P individual transposers 201-1, 201-2,. . . , 201-P. The individual transposers 201-1, 201-2, . . . , 201-Pperform a harmonic transposition of the input signal as outlined in thecontext of FIG. 1. Typically, each of the individual transposers 201-1,201-2, . . . , 201-P performs a harmonic transposition of a differenttransposition order T. By way of example, transposer 201-1 may perform atransposition of order T=1, transposer 201-2 may perform a transpositionof order T=2, . . . , and transposer 201-P may perform a transpositionof order T=P. However, in generic terms, any of the transposers 201-1, .. . , 201-P may perform a harmonic transposition of an arbitrarytransposition order T. The contributions, i.e. the output signals of theindividual transposers 201-1, 201-2, . . . , 201-P may be summed in thecombiner 202 to yield the combined transposer output.

It should be noted that each transposer 201-1, 201-2, . . . , 201-Prequires an analysis and a synthesis filter bank as depicted in FIG. 1.Moreover, the usual implementation of the individual transposers 201-1,201-2, . . . , 201-P will typically change the sampling rate of theprocessed input signal by different amounts. By way of example, thesampling rate of the output signal of the transposer 201-P may be Ttimes higher than the sampling rate of the input signal to thetransposer 201-P, wherein T is the transposition order applied by thetransposer 201-P. This may be due to a bandwidth expansion factor of Tused within the transposer 201-P, i.e. due to the use of a synthesisfilter bank which has T times more subchannels than the analysis filterbank. By doing this the sampling rate and the Nyqvist frequency isincreased by a factor T. As a consequence, the individual time domainsignals may need to be resampled in order to allow for a combining ofthe different output signals in the combiner 202. The resampling of thetime domain signals can be carried out on the input side or on theoutput side of each individual transposer 201-1, 201-2, . . . , 201-P.

FIG. 3 illustrates an exemplary configuration of a multiple harmonictransposer or multiple transposer system 300 performing several ordersof transposition and using a common analysis filter bank 301. A startingpoint for the design of the multiple transposer 300 may be to design theindividual transposers 201-1, 201-2, . . . , 201-P of FIG. 2 such thatthe analysis filter banks (reference sign 101 in FIG. 1) of alltransposers 201-1, 201-2, . . . , 201-P are identical and can bereplaced by a single analysis filter bank 301. As a consequence, thetime domain input signal is transformed into a single set of frequencydomain subband signals, i.e. a single set of analysis subband signals.These subband signals are submitted to different nonlinear processingunits 302-1, 302-2, . . . , 302-P for different orders of transposition.As outlined above in the context of FIG. 1 each nonlinear processingunit performs a modification of the phase and/or amplitude of thesubband signals and this modification differs for different orders oftransposition. Subsequently, the differently modified subband signals orsubbands have to be submitted to different synthesis filter banks 303-1,303-2, . . . , 303-P corresponding to the different nonlinear processingunits 302-1, 302-2, . . . , 302-P. As an outcome, P differentlytransposed time domain output signals are obtained which are summed inthe combiner 304 to yield the combined transposer output.

It should be noted that if the synthesis filter banks 303-1, 303-2, . .. , 303-P corresponding to the different transposition orders operate atdifferent sampling rates, e.g. by using different degrees of bandwidthexpansion, the time domain output signals of the different synthesisfilter banks 303-1, 303-2, . . . , 303-P need to be differentlyresampled in order to align the P output signals to a common time grid,prior to their summation in combiner 304.

FIG. 4 illustrates an example operation of a multiple harmonictransposer 400 using several orders of transposition, while using acommon synthesis filter bank 404. The starting point for the design ofsuch a multiple transposer 400 may be the design of the individualtransposers 201-1, 201-2, . . . , 201-P of FIG. 2 such that thesynthesis filter banks of all transposers are identical and can bereplaced by a single synthesis filter bank 404. It should be noted thatin an analogous manner as in the situation shown in FIG. 3, thenonlinear processing units 402-1, 402-2, . . . , 402-P are different foreach transposition order. Furthermore, the analysis filter banks 401-1,401-2, . . . , 401-P are different for the different transpositionorders. As such, a set of P analysis filter banks 401-1, 401-2, . . . ,401-P determines P sets of analysis subband signals. These P sets ofanalysis subband signals are submitted to corresponding nonlinearprocessing units 402-1, 402-2, . . . , 402-P to yield P sets of modifiedsubband signals. These P sets of subband signals may be combined in thefrequency domain in the combiner 403 to yield a combined set of subbandsignals as an input to the single synthesis filter bank 404. Thiscombination in combiner 403 may comprise the feeding of differentlyprocessed subband signals into different subband ranges and/or thesuperposing of contributions of subband signals to overlapping subbandranges. In other words, different analysis subband signals which havebeen processed with different transposition orders may cover overlappingfrequency ranges. By way of example, a second order transposer maytranspose the analysis subband [2A,2B] to the subband range [4A,4B]. Atthe same time, a fourth order transposer may transpose the analysissubband [A,B] to the same subband range [4A,4B]. In such cases, thesuperposing contributions may be combined, e.g. added and/or averaged,by the combiner 403. The time domain output signal of the multipletransposer 400 is obtained from the common synthesis filter bank 404. Ina similar manner as outlined above, if the analysis filter banks 401-1,401-2, . . . , 401-P operate at different sampling rates, the timedomain signals input to the different analysis filter banks 401-1,401-2, . . . , 401-P may need to be resampled in order to align theoutput signals of the different nonlinear processing units 402-1, 402-2,. . . , 402-P to the same time grid.

FIG. 5 illustrates the operation of a multiple harmonic transposer 500using several orders of transposition and comprising a single commonanalysis filter bank 501 and a single common synthesis filter bank 504.In this case, the individual transposers 201-1, 201-2, . . . , 201-P ofFIG. 2 should be designed such that both, the analysis filter banks andthe synthesis filter banks of all the P harmonic transposers areidentical. If the condition of identical analysis and synthesis filterbanks for the different P harmonic transposers is met, then theidentical filter banks can be replaced by a single analysis filter bank501 and a single synthesis filter bank 504. The advanced nonlinearprocessing units 502-1, 502-2, . . . , 502-P output differentcontributions to partly overlapping frequency ranges that are combinedin the combiner 503 to yield a combined input to the respective subbandsof the synthesis filter bank 504. Similarly to the multiple harmonictransposer 400 depicted in FIG. 4, the combination in the combiner 503may comprise the feeding of the different output signals of theplurality of nonlinear processing units 502-1, 502-2, . . . , 502-P intodifferent subband ranges, and the superposing of multiple contributingoutputs to overlapping subband ranges.

As already indicated above, the nonlinear processing 102 typicallyprovides a number of subbands at its output which corresponds to thenumber of subbands at the input. The non-linear processing 102 typicallymodifies the phase and/or the amplitude of the subband or the subbandsignal according to the underlying transposition order T. By way ofexample a subband at the input is converted to a subband at the outputwith T times higher frequency, i.e. a subband at the input to thenonlinear processing 102, i.e. the analysis subband,

$\left\lbrack {{\left( {k - \frac{1}{2}} \right)\Delta\; f},{\left( {k + \frac{1}{2}} \right)\Delta\; f}} \right\rbrack$may be transposed to a subband at the output of the nonlinear processing102, i.e. the synthesis subband,

$\left\lbrack {{\left( {k - \frac{1}{2}} \right)T\;\Delta\; f},{\left( {k + \frac{1}{2}} \right)T\;\Delta\; f}} \right\rbrack,$wherein k is a subband index number and Δf if the frequency resolutionof the analysis filter bank. In order to allow for the use of commonanalysis filter banks 501 and common synthesis filter banks 504, one ormore of the advanced processing units 502-1, 502-2, . . . , 502-P may beconfigured to provide a number of output subbands which may be differentfrom the number of input subbands.

In the following, the principles of advanced nonlinear processing in thenonlinear processing units 502-1, 502-2, . . . , 502-P will be outlined.For this purpose, it is assumed that

-   -   the analysis filter bank and the synthesis filter bank share the        same physical time stride parameter Δt.    -   the analysis filter bank has a physical frequency resolution Δf.    -   the synthesis filter bank has a physical frequency resolution        QΔf where the resolution factor Q≧1 is an integer.

Furthermore, it is assumed that the filter banks are evenly stacked,i.e. the subband with index zero is centered around the zero frequency,such that the analysis filter bank center frequencies are given by kΔfwhere the analysis subband index k=1, . . . , K_(A)−1 and K_(A) is thenumber of subbands of the analysis filter bank. The synthesis filterbank center frequencies are given by kQΔf where the synthesis subbandindex n=1, . . . , N_(S)−1 and N_(S) is the number of subbands of thesynthesis filter bank.

When performing a conventional transposition of integer order T≧1 asshown in FIG. 1, the resolution factor Q is selected as Q=T and thenonlinearly processed analysis subband k is mapped into the synthesissubband with the same index n=k. The nonlinear processing 102 typicallycomprises multiplying the phase of a subband or subband signal by thefactor T. I.e. for each sample of the filter bank subbands one may writeθ_(S)(k)=Tθ _(A)(k),  (1)where θ_(A)(k) is the phase of a (complex) sample of the analysissubband k and θ_(S)(k) is the phase of a (complex) sample of thesynthesis subband k. The magnitude or amplitude of a sample of thesubband may be kept unmodified or may be increased or decreased by aconstant gain factor. Due to the fact that T is an integer, theoperation of equation (1) is independent of the definition of the phaseangle.

In conventional multiple transposers, the resolution factor Q of ananalysis/synthesis filter bank is selected to be equal to thetransposition order T of the respective transposer, i.e. Q=T. In thiscase, the frequency resolution of the synthesis filter bank is TΔf andtherefore depends on the transposition order T. Consequently, it isnecessary to use different filter banks for different transpositionorders T either in the analysis or synthesis stage. This is due to thefact that the transposition order T defines the quotient of physicalfrequency resolutions, i.e. the quotient of the frequency resolution Δfof the analysis filter bank and the frequency resolution TΔf of thesynthesis filter bank.

In order to be able to use a common analysis filter bank 501 and acommon synthesis filter bank 504 for a plurality of differenttransposition orders T, it is proposed to set the frequency resolutionof the synthesis filter bank 504 to QΔf, i.e. it is proposed to make thefrequency resolution of the synthesis filter bank 504 independent of thetransposition order T. Then the question arises of how to implement atransposition of order T when the resolution factor Q, i.e. the quotientQ of the physical frequency resolution of the analysis and synthesisfilter bank, does not necessarily obey the relation Q=T.

As outlined above, a principle of harmonic transposition is that theinput to the synthesis filter bank subband n with center frequency nQΔfis determined from an analysis subband at a T times lower centerfrequency, i.e. at the center frequency nQΔf/T. The center frequenciesof the analysis subbands are identified through the analysis subbandindex k as kΔf. Both expressions for the center frequency of theanalysis subband index, i.e. nQΔf/T and kΔf, may be set equal. Takinginto account that the index n is an integer value, the expression

$\frac{nQ}{T}$is a rational number which can be expressed as the sum of an integeranalysis subband index k and a remainder rε{0, 1/T, 2/T, . . . ,(T−1)/T} such that

$\begin{matrix}{\frac{nQ}{T} = {k + {r.}}} & (2)\end{matrix}$

As such, it may be stipulated that the input to a synthesis subband withsynthesis subband index n may be derived, using a transposition of orderT, from the analysis subband with the index k given by equation (2). Inview of the fact that

$\frac{nQ}{T}$is a rational number, the remainder r may be unequal to 0 and the valuek+r may be greater than the analysis subband index k and smaller thanthe analysis subband index k+1, i.e. k≦k+r≦k+1. Consequently, the inputto a synthesis subband with synthesis subband index n should be derived,using a transposition of order T, from the analysis subbands with theanalysis subband index k and k+1, wherein k is given by equation (2). Inother words, the input of a synthesis subband may be derived from twoconsecutive analysis subbands.

As an outcome of the above, the advanced nonlinear processing performedin a nonlinear processing unit 502-1, 502-2, . . . , 502-P may comprisethe step of considering two neighboring analysis subbands with index kand k+1 in order to provide the output for synthesis subband n. For atransposition order T, the phase modification performed by the nonlinearprocessing unit 502-1, 502-2, . . . , 502-P may for example be definedby the linear interpolation rule,θ_(S)(n)=T(1−r)θ_(A)(k)+Trθ _(A)(k+1),  (3)where θ_(A)(k) is the phase of a sample of the analysis subband k,θ_(A)(k+1) is the phase of a sample of the analysis subband k+1, andθ_(S)(n) is the phase of a sample of the synthesis subband n. If theremainder r is close to zero, i.e. if the value k+r is close to k, thenthe main contribution of the phase of the synthesis subband sample isderived from the phase of the analysis subband sample of subband k. Onthe other hand, if the remainder r is close to one, i.e. if the valuek+r is close to k+1, then the main contribution of the phase of thesynthesis subband sample is derived from the phase of the analysissubband sample of subband k+1. It should be noted that the phasemultipliers T(1−r) and Tr are both integers such that the phasemodifications of equation (3) are well defined and independent of thedefinition of the phase angle.

Concerning the magnitudes of the subband samples, the followinggeometrical mean value may be selected for the determination of themagnitude of the synthesis subband samples,a _(S)(n)=a _(A)(k)^((1−r)) a _(A)(k+1)^(r),  (4)where a_(S)(n) denotes the magnitude of a sample of the synthesissubband n, a_(A)(k) denotes the magnitude of a sample of the analysissubband k and a_(A)(k+1) denotes the magnitude of a sample of theanalysis subband k+1. It should be noted that other interpolation rulesfor the phase and/or the magnitude may be contemplated.

For the case of an oddly stacked filter bank where the analysis filterbank center frequencies are given by

$\left( {n + \frac{1}{2}} \right)\Delta\; f$with k=1, . . . , K_(A)−1 and the synthesis filter bank centerfrequencies are given by

$\left( {n + \frac{1}{2}} \right)\frac{Q\;\Delta\; f}{T}$with n=1, . . . , N_(S)−1, an corresponding equation to equation (2) maybe derived by equating the transposed synthesis filter bank centerfrequency

$\left( {n + \frac{1}{2}} \right)\frac{Q\;\Delta\; f}{T}$and the analysis filter bank center frequency

$\left( {k + \frac{1}{2}} \right)\Delta\;{f.}$Assuming an integer index k and a remainder rε[0,1[ the followingequation for oddly stacked filter banks can be derived:

$\begin{matrix}{{\left( {n + \frac{1}{2}} \right)\frac{Q\;}{T}} = {k + \frac{1}{2} + {r.}}} & (5)\end{matrix}$

The skilled person will appreciate that if T−Q, i.e. the differencebetween the transposition order and the resolution factor, is even,T(1−r) and Tr are both integers and the interpolation rules of equations(3) and (4) can be used.

The mapping of analysis subbands into synthesis subbands is illustratedin FIG. 5b . FIG. 5b shows four diagrams for different transpositionorders T=1 to T=4. Each diagram illustrates how the source bins 510,i.e. the analysis subbands, are mapped into target bins 530, i.e.synthesis subbands. For ease of illustration, it is assumed that theresolution factor Q is equal to one. In other words, FIG. 5b illustratesthe mapping of analysis subband signals to synthesis subband signalsusing Eq. (2) and (3). In the illustrated example the analysis/synthesisfilter bank is evenly stacked, with Q=1 and the maximum transpositionorder T=4.

In the illustrated case, equation (2) may be written as

$\frac{n}{T} = {k + {r.}}$Consequently, for a transposition order T=1, an analysis subband with anindex k is mapped to a corresponding synthesis subband n and theremainder r is always zero. This can be seen in FIG. 5b where forexample source bin 511 is mapped one to one to a target bin 531.

In case of transposition order T=2, the remainder r takes on the values0 and ½ and a source bin is mapped to a plurality of target bins. Whenreversing the perspective, it may be stated that each target bin 532,535 receives a contribution from up to two source bins. This can be seenin FIG. 5b , where the target bin 535 receives a contribution fromsource bins 512 and 515. However, the target bin 532 receives acontribution from source bin 512 only. If it is assumed that target bin532 has an even index n, e.g. n=10, then equation (2) specifies thattarget bin 532 receives a contribution from the source bin 512 with anindex k=n/2, e.g. k=5. The remainder r is zero, i.e. there is nocontribution from the source bin 515 with index k+1, e.g. k+1=6. Thischanges for target bin 535 with an uneven index n, e.g. n=11. In thiscase, equation (2) specifies that target bin 535 receives contributionsfrom the source bin 512 (index k=5) and source bin 515 (index k+1=6).This applies in a similar manner to higher transposition orders T, e.g.T=3 and T=4, as shown in FIG. 5 b.

A further interpretation of the above advanced nonlinear processing maybe as follows. The advanced nonlinear processing may be understood as acombination of a transposition of a given order T into intermediatesubband signals on an intermediate frequency grid TΔf, and a subsequentmapping of the intermediate subband signals to a frequency grid definedby a common synthesis filter bank, i.e. by a frequency grid QΔf. Inorder to illustrate this interpretation, reference is made again to FIG.5b . However, for this illustration, the source bins 510 are consideredto be intermediate subbands derived from the analysis subbands using anorder of transposition T. These intermediate subbands have a frequencygrid given by TΔf. In order to generate synthesis subband signals on apre-defined frequency grid QΔf given by the target bins 530, the sourcebins 510, i.e. the intermediate subbands having the frequency grid TΔf,need to be mapped onto the pre-defined frequency grid QΔf. This can beperformed by determining a target bin 530, i.e. a synthesis subbandsignal on the frequency grid QΔf, by interpolating one or two sourcebins 510, i.e. intermediate subband signals on the frequency grid TΔf.In a preferred embodiment, linear interpolation is used, wherein theweights of the interpolation are inversely proportional to thedifference between the center frequency of the target bin 530 and thecorresponding source bin 510. By way of example, if the difference iszero, then the weight is 1, and if the difference is TΔf then the weightis 0.

In summary, a nonlinear processing method has been described whichallows the determination of contributions to a synthesis subband bymeans of transposition of several analysis subbands. The nonlinearprocessing method enables the use of single common analysis andsynthesis subband filter banks for different transposition orders,thereby significantly reducing the computational complexity of multipleharmonic transposers.

FIGS. 6a and 6b illustrate example analysis/synthesis filter banks usinga M=1024 point FFT/DFT (Fast Fourier Transform or Discrete FourierTransform) for multiple transposition orders of T=2, 3, 4. FIG. 6aillustrates the conventional case of a multiple harmonic transposer 600using a common analysis filter bank 601 and separate synthesis filterbanks 602, 603, 604 for each transposition factor T=2, 3, 4. FIG. 6ashows the analysis windows v_(A) 611 and the synthesis windows v_(S)612, 613, 614 applied at the analysis filter bank 601 and the synthesisfilter banks 602, 603, 604, respectively. In the illustrated example,the analysis window v_(A) 611 has a length L_(A)=1024 which is equal tothe size M of the FFT or DFT of the analysis/synthesis filter banks 601,602, 603, 604. In a similar manner, the synthesis windows v_(S) 612,613, 614 have a length of L_(S)=1024 which is equal to the size M of theFFT or DFT.

FIG. 6a also illustrates the hop size Δs_(A) employed by the analysisfilter bank 601 and the hop size Δs_(S) employed the synthesis filterbanks 602, 603, 604, respectively. The hop size Δs corresponds to thenumber of data samples by which the respective window 611, 612, 613, 614is moved between successive transformation steps. The hop size Δsrelates to the physical time stride Δt via the sampling rate of theunderlying signal, i.e. Δs=f_(S)Δt, wherein f_(S) is the sampling rate.

It can be seen that the analysis window 611 is moved by a hop size 621of 128 samples. The synthesis window 612 corresponding to atransposition of order T=2 is moved by a hop size 622 of 256 samples,i.e. a hop size 622 which is twice the hop size 621 of the analysiswindow 611. As outlined above, this leads to a time stretch of thesignal by the factor T=2. Alternatively, if a T=2 times higher samplingrate is assumed, the difference between the analysis hop size 621 andthe synthesis hop size 622 leads to a harmonic transposition of orderT=2. I.e. a time stretch by an order T may be converted into a harmonictransposition by performing a sampling rate conversion of order T.

In a similar manner, it can be seen that the synthesis hop size 623associated with the harmonic transposer of order T=3 is T=3 times higherthan the analysis hop size 621, and the synthesis hop size 624associated with the harmonic transposer of order T=4 is T=4 times higherthan the analysis hop size 621. In order to align the sampling rates ofthe 3^(rd) order transposer and the 4^(th) order transposer with theoutput sampling rate of the 2^(nd) order transposer, the 3^(rd) ordertransposer and the 4^(th) order transposer comprise a factor3/2—downsampler 633 and a factor 2—downsampler 634, respectively. Ingeneral terms, the T^(th) order transposer would comprise a factorT/2—downsampler, if an output sampling rate is requested, which is 2times higher than the input sampling rate. I.e. no downsampling isrequired for the harmonic transposer of order T=2.

Finally, FIG. 6a illustrates the separate phase modification units 642,643, 644 for the transposition order T=2, 3, 4, respectively. Thesephase modification units 642, 643, 644 perform a multiplication of thephase of the respective subband signals by the transposition order T=2,3, 4, respectively (see Equation (1)).

An efficient combined filter bank structure for the transposer can beobtained by limiting the multiple transposer of FIG. 6a to a singleanalysis filter bank 601 and a single synthesis filter bank 602. The3^(rd) and 4^(th) order harmonics are then produced in a non-linearprocessing unit 650 within a 2^(nd) order filter bank as depicted inFIG. 6b . FIG. 6b shows an analysis filter bank comprising a 1024 pointforward FFT unit 601 and an analysis window 611 which is applied on theinput signal x with an analysis hop size 621. The synthesis filter bankcomprises a 1024 point inverse FFT unit 602 and a synthesis window 612which is applied with a synthesis hop size 622. In the illustratedexample the synthesis hop size 622 is twice the analysis hop size 621.Furthermore, the sampling rate of the output signal y is assumed to betwice the sampling rate of the input signal x.

The analysis/synthesis filter bank of FIG. 6b comprises a singleanalysis filter bank and a single synthesis filter bank. By usingadvanced nonlinear processing 650 in accordance to the methods outlinedin the context of FIG. 5 and FIG. 5b , i.e. the advanced non-linearprocessing performed in the units 502-1, . . . , 502-P, thisanalysis/synthesis filter bank may be used to provide a multipletransposer, i.e. a harmonic transposer for a plurality of transpositionorders T.

As has been outlined in the context of FIGS. 5 and 5 b, the one-to-onemapping of analysis subbands to corresponding synthesis subbandsinvolving a multiplication of the phase of the subband signals by therespective transposition order T, may be generalized to interpolationrules (see Equations (3) and (4)) involving one or more subband signals.It has been outlined that if the physical spacing QΔf of the synthesisfilter bank subbands is Q times the physical spacing Δf of the analysisfilter bank, the input to the synthesis band with index n is obtainedfrom the analysis bands with indices k and k+1. The relationship betweenthe indexes n and k is given by Equation (2) or (5), depending onwhether the filter banks are evenly or unevenly stacked. A geometricalinterpolation for the magnitudes is applied with powers 1−r and r(Equation (4)) and the phases are linearly combined with weights T(1−r)and Tr (Equation (3)). For the illustrated case where Q=2, the phasemappings for each transposition factor are illustrated graphically inFIG. 7.

In a similar manner to the case of Q=1 illustrated in FIG. 5, a targetsubband or target bin 730 receives contributions from up to two sourcesubbands or source bins 710. In the case T=Q=2, each phase modifiedsource bin 711 is assigned to a corresponding target bin 731. For highertransposition orders T>Q, a target bin 735 may be obtained from onecorresponding phase modified source bin 715. This is the case if theremainder r obtained from Equation (2) or (5) is zero. Otherwise, atarget bin 732 is obtained by interpolating two phase modified sourcebins 712 and 715.

The above mentioned non-linear processing is performed in the multipletransposer unit 650 which determines target bins 730 for the differentorders of transposition T=2, 3, 4 using advanced non-linear processingunits 502-2, 502-3, 502-4. Subsequently, corresponding target bins 730are combined in a combiner unit 503 to yield a single set of synthesissubband signals which are fed to the synthesis filter bank. As outlinedabove, the combiner unit 503 is configured to combine a plurality ofcontributions in overlapping frequency ranges from the output of thedifferent non-linear processing units 502-2, 502-3, 502-4.

In the following, the harmonic transposition of transient signals usingharmonic transposers is outlined. In this context, it should be notedthat harmonic transposition of order T using analysis/synthesis filterbanks may be interpreted as time stretching of an underlying signal byan integer transposition factor T followed by a downsampling and/orsample rate conversion. The time stretching is performed such thatfrequencies of sinusoids which compose the input signal are maintained.Such time stretching may be performed using the analysis/synthesisfilter bank in combination with intermediate modification of the phasesof the subband signals based on the transposition order T. As outlinedabove, the analysis filter bank may be a windowed DFT filter bank withanalysis window v_(A) and the synthesis filter bank may be a windowedinverse DFT filter bank with synthesis window v_(S). Suchanalysis/synthesis transform is also referred to as short-time FourierTransform (STFT).

A short-time Fourier transform is performed on a time-domain inputsignal x to obtain a succession of overlapped spectral frames. In orderto minimize possible side-band effects, appropriate analysis/synthesiswindows, e.g. Gaussian windows, cosine windows, Hamming windows, Hannwindows, rectangular windows, Bartlett windows, Blackman windows, andothers, should be selected. The time delay at which every spectral frameis picked up from the input signal x is referred to as the hop size Δsor physical time stride Δt. The STFT of the input signal x is referredto as the analysis stage and leads to a frequency domain representationof the input signal x. The frequency domain representation comprises aplurality of subband signals, wherein each subband signal represents acertain frequency component of the input signal.

For the purpose of time-stretching of the input signal, each subbandsignal may be time-stretched, e.g. by delaying the subband signalsamples. This may be achieved by using a synthesis hop-size which isgreater than the analysis hop-size. The time domain signal may berebuilt by performing an inverse (Fast) Fourier transform on all framesfollowed by a successive accumulation of the frames. This operation ofthe synthesis stage is referred to as overlap-add operation. Theresulting output signal is a time-stretched version of the input signalcomprising the same frequency components as the input signal. In otherwords, the resulting output signal has the same spectral composition asthe input signal, but it is slower than the input signal i.e. itsprogression is stretched in time.

The transposition to higher frequencies may then be obtainedsubsequently, or in an integrated manner, through downsampling of thestretched signals or by performing a sample-rate conversion of the timestretched output signal. As a result the transposed signal has thelength in time of the initial signal, but comprises frequency componentswhich are shifted upwards by a pre-defined transposition factor.

In view of the above, the harmonic transposition of transient signalsusing harmonic transposers is described by considering as a startingpoint the time stretching of a prototype transient signal, i.e. adiscrete time Dirac pulse at time instant t=t₀,

${\delta\left( {t - t_{0}} \right)} = \left\{ {\begin{matrix}{1,{t = t_{0}}} \\{0,{t \neq t_{0}}}\end{matrix}.} \right.$

The Fourier transform of such a Dirac pulse has unit magnitude and alinear phase with a slope proportional to t₀:

${{X\left( \Omega_{m} \right)} = {{\sum\limits_{n = {- \infty}}^{\infty}{{\delta\left( {n - t_{0}} \right)}{\exp\left( {{- {j\Omega}_{m}}n} \right)}}} = {\exp\left( {{- {j\Omega}_{m}}t_{0}} \right)}}},$wherein

$\Omega_{m} = {2\pi\frac{m}{M}}$is the center frequency of the m^(th) subband signal of the STFTanalysis and M is the size of the discrete Fourier transform (DFT). SuchFourier transform can be considered as the analysis stage of theanalysis filter bank described above, wherein a flat analysis windowv_(A) of infinite duration is used. In order to generate an outputsignal y which is time-stretched by a factor T, i.e. a Dirac pulseδ(t−Tt₀) at the time instant t=Tt₀, the phase of the analysis subbandsignals should be multiplied by the factor T in order to obtain thesynthesis subband signal Y(Ω_(m))=exp(−jΩ_(m)Tt₀) which yields thedesired Dirac pulse δ(t−Tt₀) as an output of an inverse FourierTransform.

However, it should be noted that the above considerations refer to ananalysis/synthesis stage using analysis and synthesis windows ofinfinite lengths. Indeed, a theoretical transposer with a window ofinfinite duration would give the correct stretch of a Dirac pulseδ(t−t₀). For a finite duration windowed analysis, the situation isscrambled by the fact that each analysis block is to be interpreted asone period interval of a periodic signal with a period equal to the sizeof the DFT.

This is illustrated in FIG. 8 which shows the analysis and synthesis 800of a Dirac pulse δ(t−t₀). The upper part of FIG. 8 shows the input tothe analysis stage 810 and the lower part of FIG. 8 shows the output ofthe synthesis stage 820. The upper and lower graphs represent the timedomain. The stylized analysis window 811 and synthesis window 821 aredepicted as triangular (Bartlett) windows. The input pulse δ(t−t₀) 812at time instant t=t₀ is depicted on the top graph 810 as a verticalarrow. It is assumed that the DFT transform block is of sizeM=L=L_(A)=L_(S), i.e. the size of the DFT transform is chosen to beequal to the size of the windows. The phase multiplication of thesubband signals by the factor T will produce the DFT analysis of a Diracpulse δ(t−Tt₀) at t=Tt₀, however, of a Dirac pulse periodized to a Diracpulse train with period L. This is due to the finite length of theapplied window and Fourier Transform. The periodized pulse train withperiod L is depicted by the dashed arrows 823, 824 on the lower graph.

In a real-world system, the pulse train actually contains a few pulsesonly (depending on the transposition factor), one main pulse, i.e. thewanted term, a few pre-pulses and a few post-pulses, i.e. the unwantedterms. The pre- and post-pulses emerge because the DFT is periodic (withL). When a pulse is located within an analysis window, so that thecomplex phase gets wrapped when multiplied by T (i.e. the pulse isshifted outside the end of the window and wraps back to the beginning),an unwanted pulse emerges within the synthesis window. The unwantedpulses may have, or may not have, the same polarity as the input pulse,depending on the location in the analysis window and the transpositionfactor.

In the example of FIG. 8, the synthesis windowing uses a finite windowv_(S) 821. The finite synthesis window 821 picks the desired pulseδ(t−Tt₀) at t=Tt₀ which is depicted as a solid arrow 822 and cancels theother unwanted contributions which are shown as dashed arrows 823, 824.

As the analysis and synthesis stage move along the time axis accordingto the hop factor Δs or the time stride Δt, the pulse δ(t−t₀) 812 willhave another position relative to the center of the respective analysiswindow 811. As outlined above, the operation to achieve time-stretchingconsists in moving the pulse 812 to T times its position relative to thecenter of the window. As long as this position is within the window 821,this time-stretch operation guarantees that all contributions add up toa single time stretched synthesized pulse δ(t−Tt₀) at t=Tt₀.

However, a problem occurs for the situation of FIG. 9, where the pulseδ(t−t₀) 912 moves further out towards the edge of the DFT block. FIG. 9illustrates a similar analysis/synthesis configuration 900 as FIG. 8.The upper graph 910 shows the input to the analysis stage and theanalysis window 911, and the lower graph 920 illustrates the output ofthe synthesis stage and the synthesis window 921. When time-stretchingthe input Dirac pulse 912 by a factor T, the time stretched Dirac pulse922, i.e. δ(t−Tt₀), comes to lie outside the synthesis window 921. Δtthe same time, another Dirac pulse 924 of the pulse train, i.e.δ(t−Tt₀+L) at time instant t=Tt₀−L, is picked up by the synthesiswindow. In other words, the input Dirac pulse 912 is not delayed to a Ttimes later time instant, but it is moved forward to a time instant thatlies before the input Dirac pulse 912. The final effect on the audiosignal is the occurrence of a pre-echo at a time distance of the scaleof the rather long transposer windows, i.e. at a time instant t=Tt₀−Lwhich is L−(T−1)t₀ earlier than the input Dirac pulse 912.

The principle of the solution to this problem is described in referenceto FIG. 10. FIG. 10 illustrates an analysis/synthesis scenario 1000similar to FIG. 9. The upper graph 1010 shows the input to the analysisstage with the analysis window 1011, and the lower graph 1020 shows theoutput of the synthesis stage with the synthesis window 1021. The DFTsize is adapted so as to avoid pre-echoes. This may be achieved bysetting the size M of the DFT such that no unwanted Dirac pulse imagesfrom the resulting pulse train are picked up by the synthesis window.The size of the DFT transform 1001 is increased to M=FL, where L is thelength of the window function 1002 and the factor F is a frequencydomain oversampling factor. In other words, the size of the DFTtransform 1001 is selected to be larger than the window size 1002. Inparticular, the size of the DFT transform 1001 may be selected to belarger than the window size 1002 of the synthesis window. Due to theincreased length 1001 of the DFT transform, the period of the pulsetrain comprising the Dirac pulses 1022, 1024 is FL. By selecting asufficiently large value of F, i.e. by selecting a sufficiently largefrequency domain oversampling factor, undesired contributions to thepulse stretch can be cancelled. This is shown in FIG. 10, where theDirac pulse 1024 at time instant t=Tt₀−FL lies outside the synthesiswindow 1021. Therefore, the Dirac pulse 1024 is not picked up by thesynthesis window 1021 and by consequence, pre-echoes can be avoided.

It should be noted that in a preferred embodiment the synthesis windowand the analysis window have equal “nominal” lengths (measured in thenumber of samples). However, when using implicit resampling of theoutput signal by discarding or inserting samples in the frequency bandsof the transform or filter bank, the synthesis window size (measured inthe number of samples) will typically be different from the analysissize, depending on the resampling and/or transposition factor.

The minimum value of F, i.e. the minimum frequency domain oversamplingfactor, can be deduced from FIG. 10. The condition for not picking upundesired Dirac pulse images may be formulated as follows: For any inputpulse δ(t−t₀) at position

${t = {t_{0} < \frac{L}{2}}},$i.e. for any input pulse comprised within the analysis window 1011, theundesired image δ(t−Tt₀+FL) at time instant t=Tt₀−FL must be located tothe left of the left edge of the synthesis window at

$t = {- {\frac{L}{2}.}}$In an equivalent manner, the condition

${{T\frac{L}{2}} - {FL}} \leq {- \frac{L}{2}}$must be met, which leads to the rule

$\begin{matrix}{F \geq {\frac{T + 1}{2}.}} & (6)\end{matrix}$

As can be seen from formula (6), the minimum frequency domainoversampling factor F is a function of the transposition order T. Morespecifically, the minimum frequency domain oversampling factor F isproportional to the transposition order T.

By repeating the line of thinking above for the case where the analysisand synthesis windows have different lengths one obtains a more generalformula. Let L_(A) and L_(S) be the lengths of the analysis andsynthesis windows (measured in the number of samples), respectively, andlet M be the DFT size employed. The general rule extending formula (6)is then

$\begin{matrix}{M \geq {\frac{{TL}_{A} + L_{S}}{2}.}} & (7)\end{matrix}$

That this rule indeed is an extension of (6) can be verified byinserting M=FL, and L_(A)=L_(S)=L in (7) and dividing by L on both sideof the resulting equation.

The above analysis is performed for a rather special model of atransient, i.e. a Dirac pulse. However, the reasoning can be extended toshow that when using the above described time-stretching and/or harmonictransposition scheme, input signals which have a near flat spectralenvelope and which vanish outside a time interval [a,b] will bestretched to output signals which are small outside the interval[Ta,Tb]. It can also be verified, by studying spectrograms of real audioand/or speech signals, that pre-echoes disappear in the stretched ortransposed signals when the above described rule for selecting anappropriate frequency domain oversampling factor is respected. A morequantitative analysis also reveals that pre-echoes are still reducedwhen using frequency domain oversampling factors which are slightlyinferior to the value imposed by the condition of formula (6) or (7).This is due to the fact that typical window functions v_(S) are smallnear their edges, thereby attenuating undesired pre-echoes which arepositioned near the edges of the window functions.

In summary, a way to improve the transient response of frequency domainharmonic transposers, or time-stretchers, has been described byintroducing an oversampled transform, where the amount of oversamplingis a function of the transposition factor chosen. The improved transientresponse of the transposer is obtained by means of frequency domainoversampling.

In the multiple transposer of FIG. 6, frequency domain oversampling maybe implemented by using DFT kernels 601, 602, 603, 604 of length 1024Fand by zero padding the analysis and synthesis windows symmetrically tothat length. It should be noted that for complexity reasons, it isbeneficial to keep the amount of oversampling low. If formula (6) isapplied to the multiple transposer of FIG. 6, an oversampling factorF=2.5 should be applied to cover all the transposition factors T=2, 3,4. However, it can be shown that the use of F=2.0 already leads to asignificant quality improvement for real audio signals.

In the following, the use of frequency domain oversampling in thecontext of combined analysis/synthesis filter banks, such as describedin the context of FIG. 5 or FIG. 6b , is described.

In general, for a combined transposition filter bank where the physicalspacing QΔf of the synthesis filter bank subbands is Q times thephysical spacing Δf of the analysis filter bank and where the physicalanalysis window duration D_(A) (measured in units of time, e.g. seconds)is also Q times that of the synthesis filter bank, D_(A)=QD_(S), theanalysis for a Dirac pulse as above will apply for all transpositionfactors T=Q, Q+1, Q+2, . . . as if T=Q. In other words, the rule for thedegree of frequency domain oversampling required in a combinedtransposition filter bank is given by

$\begin{matrix}{F \geq {\frac{Q + 1}{2}.}} & \left( {6b} \right)\end{matrix}$

In particular, it should be noted that for T>Q, the frequency domainoversampling factor

$F < \frac{T + 1}{2}$is sufficient, while still ensuring the suppression of artifacts ontransient signals caused by harmonic transposition of order T. I.e.using the above oversampling rules for the combined filter bank, it canbe seen that even when using higher transposition orders T>Q, it is notrequired to further increase the oversampling factor F. As indicated byequation (6b), it is sufficient in the combined filter bankimplementation of FIG. 6b to use an oversampling factor F=1.5 in orderto avoid the occurrence of pre-echoes. This value is lower than theoversampling factor F=2.5 required for the multiple transposer of FIG.6. Consequently, the complexity of performing frequency domainoversampling in order to improve the transient performance of multipleharmonic transposers can be further reduced when using a combinedanalysis/synthesis filter bank (instead of separate analysis and/orsynthesis filter banks for the different transposition orders).

In a more general scenario, the physical time durations of the analysisand synthesis windows D_(A) and D_(S), respectively, may be arbitrarilyselected. Then the physical spacing Δf of the analysis filter banksubbands should satisfy

$\begin{matrix}{{{\Delta\; f} \leq \frac{2}{Q\left( {D_{A} + D_{S}} \right)}},} & \left( {7b} \right)\end{matrix}$in order to avoid the described artifacts caused by harmonictransposition. It should be noted that the duration of a window Dtypically differs from the length of a window L. Whereas the length of awindow L corresponds to the number of signal samples covered by thewindow, the duration of the window D corresponds to the time interval ofthe signal covered by the window. As illustrated in FIG. 6a , thewindows 611, 612, 613, 614 have an equal length of L=1024 samples.However, the duration D_(A) of the analysis window 611 is T times theduration D_(S) of the synthesis window 612, 613, 614, wherein T is therespective transposition order and the resolution factor of therespective synthesis filter bank. In a similar manner, the durationD_(A) of the analysis window 611 in FIG. 6b is Q times the durationD_(S) of the synthesis window 612, wherein Q is the resolution factor ofthe synthesis filter bank. The duration of a window D is related to thelength of the window L via the sampling frequency f_(S), i.e. notably

$D = {\frac{L}{f_{s}}.}$In a similar manner, the frequency resolution of a transform Δf isrelated to the number of points or length M of the transform via thesampling frequency f_(S), i.e. notably

${\Delta\; f} = {\frac{f_{s}}{M}.}$Furthermore, the physical time stride Δt of a filter bank is related tothe hop size Δs of the filter bank via the sampling frequency f_(S),i.e. notably

${\Delta\; t} = {\frac{\Delta\; s}{f_{s}}.}$

Using the above relations, equation (6b) may be written as

$\begin{matrix}{{{\Delta\;{fD}_{A}} = {{Q\;\Delta\;{fD}_{s}} \leq \frac{2}{Q + 1}}},} & \left( {6c} \right)\end{matrix}$i.e. the product of the frequency resolution and the window length ofthe analysis filter bank and/or the frequency resolution and the windowlength of the synthesis filter bank should be selected to be smaller orequal to

$\frac{2}{Q + 1}.$For T>Q, the product ΔfD_(A) and/or QΔfD_(S) may be selected to begreater than

$\frac{2}{T + 1},$thereby reducing the computational complexity of the filter banks.

In the present document, various methods for performing harmonictransposition of signals, preferably audio and/or speech signals, havebeen described. Particular emphasis has been put on the computationalcomplexity of multiple harmonic transposers. In this context, a multipletransposer has been described, which is configured to perform multipleorders of transposition using a combined analysis/synthesis filter bank,i.e. a filter bank comprising a single analysis filter bank and a singlesynthesis filter bank. A multiple tranposer using a combinedanalysis/synthesis filter bank has reduced computational complexitycompared to a conventional multiple transposer. Furthermore, frequencydomain oversampling has been described in the context of combinedanalysis/synthesis filter banks. Frequency domain oversampling may beused to reduce or remove artifacts caused on transient signals byharmonic transposition. It has been shown that frequency domainoversampling can be implemented at reduced computational complexitywithin combined analysis/synthesis filter banks, compared toconventional multiple transposer implementations.

While specific embodiments of the present invention and applications ofthe invention have been described herein, it will be apparent to thoseof ordinary skill in the art that many variations on the embodiments andapplications described herein are possible without departing from thescope of the invention described and claimed herein. It should beunderstood that while certain forms of the invention have been shown anddescribed, the invention is not to be limited to the specificembodiments described and shown or the specific methods described.

The methods and systems described in the present document may beimplemented as software, firmware and/or hardware. Certain componentsmay e.g. be implemented as software running on a digital signalprocessor or microprocessor. Other components may e.g. be implemented ashardware and or as application specific integrated circuits. The signalsencountered in the described methods and systems may be stored on mediasuch as random access memory or optical storage media. They may betransferred via networks, such as radio networks, satellite networks,wireless networks or wireline networks, e.g. the internet. Typicaldevices making use of the methods described in the present document arefor example media players or setup boxes which decode audio signals. Onthe encoding side, the systems and methods may be used e.g. inbroadcasting stations and at multimedia production sites.

The invention claimed is:
 1. A system to generate a high frequencycomponent of an audio signal, comprising: an analysis filter bank havinga frequency resolution of Δf and an analysis window of duration D_(A),the analysis filter bank providing a plurality of analysis subbandsignals from the audio signal; a nonlinear processing unit determining asynthesis subband signal based on at least some of the plurality ofanalysis subband signals, wherein the at least some of the plurality ofanalysis subband signals are phase shifted by a transposition order T;and a synthesis filter bank having a frequency resolution of Q·Δf, thesynthesis filter bank generating the high frequency component from thesynthesis subband signal, wherein the value of the product of thefrequency resolution Δf and the duration D_(A) of the analysis filterbank is determined based on the frequency resolution Q·Δf of thesynthesis filter bank.
 2. The system of claim 1, wherein${{\Delta\; f} \leq \frac{2}{Q\left( {D_{A} + D_{S}} \right)}},$ whereinthe synthesis filter bank has a synthesis window of duration D_(S). 3.The system of claim 1, wherein the nonlinear processing unit determinesthe synthesis subband signal from two consecutive analysis subbandsignals.
 4. The system of claim 1, wherein the value of the productΔf·D_(A) satisfies:$\frac{2}{T + 1} < {\Delta\;{f \cdot D_{A}}} \leq {\frac{2}{Q + 1}.}$ 5.The system of claim 1, wherein the system comprises only a singleanalysis filter bank and a single synthesis filter bank for generatingthe high frequency component.
 6. The system of claim 1, wherein thevalue of the product ΔfD_(A) of the analysis filter bank equals thevalue of the product QΔfD_(S) of the synthesis filter bank, wherein thesynthesis filter bank has a synthesis window of duration D_(S).
 7. Thesystem of claim 1, wherein the analysis filter bank comprises ananalysis transformation unit to perform one of a Fourier Transform, aFast Fourier Transform, a Discrete Fourier Transform, a WaveletTransform; and the synthesis filter bank comprises a synthesistransformation unit to perform the corresponding inverse transform. 8.The system of claim 1, wherein the analysis and/or synthesis window isone of: Gaussian window; cosine window; Hamming window; Hann window;rectangular window; Bartlett window; Blackman window.
 9. The system ofclaim 1, further comprising: a second nonlinear processing unitconfigured to determine a second set of synthesis subband signals fromthe set of analysis subband signals using a second transposition orderT₂; wherein the second set of synthesis subband signals is determinedbased on a portion of the set of analysis subband signals, phase shiftedby the second transposition order T₂; wherein the transposition order Tand the second transposition order T₂ are different; and a combiningunit configured to combine the set of synthesis subband signals and thesecond set of synthesis subband signals; thereby yielding a combined setof synthesis subband signals as an input to the synthesis filter bank.10. The system of claim 9, wherein the combining unit is configured tosuperpose synthesis subband signals of the set of synthesis subbandsignals and the second set of synthesis subband signals corresponding tooverlapping frequency ranges.
 11. The system of claim 1, wherein theanalysis filter bank has a number K_(A) of analysis subbands, withK_(A)>1, where k is an analysis subband index with k=0, . . . , K_(A)−1;and the synthesis filter bank has a number N_(S) of synthesis subbands,with N_(S)>0, where n is a synthesis subband index with n=0, . . . ,N_(S)−1.
 12. The system of claim 11, wherein the nonlinear processingunit is configured to determine an n^(th) synthesis subband signal ofthe set of synthesis subband signals from a k^(th) analysis subbandsignal and a (k+1)^(th) analysis subband signal of the set of analysissubband signals.
 13. The system of claim 12, wherein the nonlinearprocessing unit is configured to determine a phase of the n^(th)synthesis subband signal as the sum of a shifted phase of the k^(th)analysis subband signal and a shifted phase of the (k+1)^(th) analysissubband signal; and/or determine a magnitude of the n^(th) synthesissubband signal as the product of an exponentiated magnitude of thek^(th) analysis subband signal and an exponentiated magnitude of the(k+1)^(th) analysis subband signal.
 14. The system of claim 13, whereinthe analysis subband index k of the analysis subband signal contributingto the synthesis subband with synthesis subband index n is given by theinteger obtained by truncating the expression ${\frac{Q}{T}n};$ whereina remainder r is given by ${\frac{Q}{T}n} - {k.}$
 15. The system ofclaim 14, wherein the nonlinear processing unit is configured todetermine the phase of the n^(th) synthesis subband signal as the sum ofthe phase of the k^(th) analysis subband signal multiplied by T(1−r) andthe phase of the (k+1)^(th) analysis subband signal multiplied by T(r);and/or determine the magnitude of the n^(th) synthesis subband signal asthe product of the magnitude of the k^(th) analysis subband signalraised to the power of (1−r) and the magnitude of the (k+1)^(th)analysis subband signal raised to the power of r.
 16. The system ofclaim 1, wherein the analysis filter bank and the synthesis filter bankare evenly stacked such that a center frequency of an analysis subbandis given by kΔf and a center frequency of a synthesis subband is givenby nQΔf.
 17. The system of claim 1, wherein the analysis filter bank andthe synthesis filter bank are oddly stacked such that a center frequencyof an analysis subband is given by$\left( {k + \frac{1}{2}} \right)\Delta\; f$ and a center frequency of asynthesis subband is given by${\left( {n + \frac{1}{2}} \right)Q\;\Delta\; f};$ and the differencebetween the transposition order T and the resolution factor Q is even.18. The system of claim 1, wherein a sampling rate of the audio signalis f_(A); the analysis transformation unit is a discrete M pointtransformation; the analysis window has a length of L_(A) samples; andthe analysis window is shifted by an analysis hop size of Δs_(A) samplesalong the audio signal; the frequency resolution is${{\Delta\; f} = \frac{f_{A}}{M}};$ the duration is${D_{A} = \frac{L_{A}}{f_{A}}};$ a physical time stride of the analysisfilter bank is ${{\Delta\; t_{A}} = \frac{\Delta\; s_{A}}{f_{A}}};$ asampling rate of the high frequency component is f_(S)=Qf_(A); thesynthesis transformation unit is a discrete M point transformation; thesynthesis window has a length of L_(S) samples; and the synthesis windowis shifted by a synthesis hop size of Δs_(S) samples along the highfrequency component; and the frequency resolution is${{Q\;\Delta\; f} = \frac{f_{s}}{M}};$ the duration is${D_{s} = \frac{L_{s}}{f_{s}}};$ a physical time stride of the synthesisfilter bank is${\Delta\; t_{s}} = {\frac{\Delta\; s_{s}}{f_{s}} = {\frac{\Delta\; s_{A}}{f_{A}} = {\Delta\;{t_{A}.}}}}$19. A method for generating a high frequency component of an audiosignal, the method comprising: applying an analysis filter bank having afrequency resolution of Δf and an analysis window of duration D_(A) onthe audio signal; determining a synthesis subband signal based on atleast two analysis subband signals, wherein the analysis subband signalsare phase shifted by a transposition order T; and applying a synthesisfilter bank having a frequency resolution of Q·Δf and a synthesis windowof duration D_(S) on the synthesis subband signal; wherein the value ofthe product of the frequency resolution Δf and the duration D_(A) of theanalysis filter bank is determined based on the frequency resolution ofthe synthesis filter bank.
 20. A method for designing a harmonictransposer to generate a high frequency component of an audio signal,the method comprising: providing an analysis filter bank having afrequency resolution of Δf and an analysis window of duration D_(A);providing a nonlinear processing unit to determine a synthesis subbandsignal based on at least two analysis subband signals, wherein theanalysis subband signals are phase shifted by a transposition order T;providing a synthesis filter bank having a frequency resolution of Q·Δfand a synthesis window of duration D_(S); and selecting the value of theproduct of the frequency resolution Δf and the duration D_(A) of theanalysis filter bank based on the frequency resolution of the synthesisfilter bank.